The present invention relates generally to electro-optic modulators and, more particularly, to optical modulators having an internal structure for matching impedance with coaxial connectors.
The operation of electro-optic modulators is based on the interaction between an electrical microwave, or radio frequency (RF), modulating signal and an optical signal. An optical modulator is typically obtained by utilizing the electro-optical effect of the modulator's waveguide material. This effect comprises changing, through an applied electric field, the index of refraction of the optical waveguide in which the optical signal propagates. This variation in time of the refractive index produces a desired phase modulation of the optical signal traveling through the waveguide. An amplitude modulator can be made by exploiting the above phase modulation in at least one arm of a waveguide interferometer, e.g., a Mach-Zehnder interferometer.
To obtain a modulator, it is necessary to have an optical waveguide carrying an optical signal and an electrode structure responsive to an applied RF signal that permits generation of the electric field necessary for modulating the optical signal. To increase the modulating effect, that is, the phase variation of the optical signal versus the amplitude of the applied RF signal, interaction between the optical signal and the electric field should be distributed along a planar microwave waveguide structure.
The optical beam is made to propagate parallel to the planar microwave waveguide structure. In this way, the optical signal undergoes phase variations induced by the microwave signal along the entire length microwave waveguide.
An example of such a substrate and electrode structure is shown in FIG. 3. FIG. 3 is a top view of an electro-optic substrate 2 with an optical waveguide 1 running through it. Electrode 3 generates an electric field along its entire length. Thus, an optical signal propagating through waveguide 1 undergoes phase modulation along the entire length of electrode 3.
To obtain an increased modulating effect, proper electro-optic substrates, wherein the applied RF field can induce a significant variation of the refraction index, are exploited to guide the optical signal. An example of a useful material for such a substrate is Lithium Niobate, LiNbO3. A known, less preferred alternative material is Lithium Tantalate, LiTaO3.
Moreover, the coupling between the propagating optical and microwave signal must be synchronous to allow the phase variation induced by the microwave signal to increasingly add up throughout the whole structure. Synchronous coupling can be achieved by proper design of the microwave line, i.e. by making the effective index of the line equal to the optical effective index. This result may be obtained in several ways, for instance by increasing the electrode thickness and growing the electrodes on a thin, low dielectric constant, buffer layer.
The optimization of the electrode region with respect to synchronous propagation and maximum electro-optical interaction usually leads to lines of very small width, which cannot be directly connected to a planar-to-coaxial transition. This problem is usually solved by means of a transmission line taper, such as 5 in FIG. 3. The taper leads at a constant characteristic impedance from the small modulator line 3 to the comparatively large dimension at the exterior of modulator 2. Standard coaxial transitions require this larger dimension as an interface. The resulting input impedance levels of the modulator, however, tend to be much lower than 50 ohms, which is the standard reference impedance for which coaxial connectors and RF generators are currently designed.
This mismatch in source and load impedances results in numerous problems, such that the source and load impedances should be “matched.” Impedance matching, as generally understood, comprises making a source impedance and a load impedance substantially equal, for instance, to allow the maximum transfer of electrical power from the source to the load. In the instant implementation, the source is an RF generator/coaxial cable, and the load is the optical modulator electrodes.
Input impedance matching is desirable in optical modulators, because besides increasing the input electrical power fed into the modulator, it also decreases multiple reflections and signal distortion. Because a change in refractive index in the substrate is directly related to the amount of RF electrical power input to the modulating electrode, the amplitude of optical modulation achievable at a given RF generator power is also increased when impedances are matched.
Patents in the field of electro-optic modulation describe various schemes, including providing an external matching network, for matching the impedance of optical modulators with their respective modulating signal sources.
U.S. Pat. No. 5,189,547 (Day et al.) describes a tunable adaptive external circuit connected to a bulk electro-optical modulator for impedance matching. This external driving circuit is connected between the signal generator and the modulator.
The driving circuit includes discrete components that are hand-adjustable to match the impedance of the modulator with that of the signal generator.
U.S. Pat. No. 5,572,610 (Toyohara) describes an impedance matching means for matching an impedance of a control signal source and a signal electrode for a wide band waveguide-type optical device.
In the field of microwaves, a “resonant” line is a line connected to a load having a drastically different impedance from the characteristic impedance of the line itself. In electro-optical modulators, the characteristic impedance of the “hot” (i.e., carrying the RF signal) electrode is typically several tens of ohms, for example 20-50 ohms. Typical configurations for a resonant modulator are: an open circuit RF electrode (“infinity” impedance of the load); and the RF electrode short-circuited to ground (“zero” impedance of the load). Other configurations are possible, as a RF electrode connected to a load having an impedance of few ohms or of several kilo-ohms, for example. A good parameter which can be used in defining “resonant” is the modulus of the F coefficient, which is defined as:   Γ  =                    Z        L            -              Z        0                            Z        L            -              Z        0            where: ZL is the impedance of the load, and                Z0 is the characteristic impedance of the line (RF electrode).        
|Γ| has a value in the range from 0 to 1. If, |Γ|=0, i.e., if ZL=Z0, the line is under a traveling-wave condition. If |Γ|˜1, i.e., if ZL=0 or ZL>>Z0, the resonance condition is met. Henceforth, the following practical definition of resonance will be used: a modulator is of the “resonant” type if |Γ|>0.5. A preferred resonance condition corresponds to |Γ|>0.8.
Resonant modulators are highly efficient in narrow bands around some resonance frequencies fo. Such high efficiency has been verified for frequencies around above some GHz, generally from 0.5 to 5 GHz, and preferably from 1 to 4 GHz. A typical frequency band of interest for resonant modulation is that around 2 GHz. An exemplary application of resonant modulators is phase modulation at 2 GHz for stimulated Brillouin scattering (SBS) suppression in cable television (CATV) systems. In such systems, high modulation efficiency can be exploited to save modulation power, resulting in less heating and reduced thermal stabilization problems.
For further details regarding resonant configuration and phase modulation for SBS suppression, please refer to WO 99/09451.
Impedance mismatch between the load and the line becomes a serious problem if the modulator design is of the resonant type. This problem gets worse the closer the |Γ| value is to 1 near the resonant frequencies of interest. In this case, namely, the center band impedance becomes almost imaginary. Applicants have determined that the impedance of a resonant modulator, e.g. the modulator of FIG. 3, has typically a real (resistive) part lower than 10 ohms and an imaginary (reactive) part higher than 50 ohms. The impedance mismatch between the RF signal source and the modulator is conventionally eliminated by attaching to an external, concentrated network.
U.S. Pat. No. 4,372,643 (Liu et al.) discloses a standing-wave, velocity matched gate including an optical directional coupler that has a pair of electrodes located over the waveguide. The electrodes form an electrical transmission line that is energized at its input by a signal source having an output impedance R. In one embodiment, the transmission line is terminated by a short circuit and the electrodes are proportioned such that the input impedance of the line has a real part that is equal to R. The imaginary component of the transmission line is resonated by an external impedance connected across the input end of the line.
U.S. Pat. No. 4,850,667 (Djupsjobacka) relates to an electrode arrangement for optoelectronic devices. A first elongate electrode has a connecting conductor for an incoming microwave signal with the aid of which a light wave is to be modulated. The connecting conductor divides the first electrode into a standing wave guide and a traveling wave guide, which is connected via a resistor to a U-shaped second electrode. It is stated in the '667 patent that the incoming modulating microwave has maximum modulating ability in the standing waveguide if its frequency is in agreement with the resonance frequency fc of the standing wave guide. In one embodiment, the connecting conductor is grounded.
U.S. Pat. No. 5,005,932 (Schaffner et al.) describes a traveling-wave electro-optical modulator with a periodic electrode structure of the intermittent interaction type. This electrode structure has a plurality of middle stubs to maintain the phase of the RF drive frequency in phase with the optical signal. This electrode structure makes possible the modulation of optical signals by RF signals above microwave frequencies. Impedance transforming and impedance matching characteristics are built into the modulator and this facilitates connection of the RF source since no extra impedance matching circuitry is required. Impedance transforming is performed by tapered input and output openings. The impedance matching is carried out by end stubs which are shorter than the middle stubs. The impedance matching stubs serve to transform the impedance of the periodic electrode structure to the impedance of an unperturbed linear RF coplanar waveguide. The impedance transforming sections serve to bring the impedance level seen by the RF signal at a location just outside the impedance matching stub up to the impedance level of the source and the load.
Applicants remark that impedance transforming circuitry and impedance matching features like those disclosed in the Schaffler et al. patent in combination with a traveling-wave relatively-broad-band electrode structure, cannot be used with a resonant modulator, as they would not allow to compensate the almost imaginary impedance of a resonant modulator.
Applicants have noticed that an external impedance matching network imposes additional costs, not only for the matching components, but also for the separate packaging and connectors for the interface between the modulator and the RF generator. Similarly, Applicants have recognized that the overall dimension of the system having an external impedance matching network is large due to these external components. As well, the reliability and repeatability of the external matching is undesirably low, due to variation in the component values and parasitic impedances of the external components.